The present invention relates to a switching power supply for supplying a stabilized DC voltage to industrial or consumer electronics equipments.
A power supply for a television set, a VTR, a personal computer, and so on, is required to stably supply a steady DC voltage. A switching power supply is preferable as such a power supply. The switching power supply uses semiconductor devices, e.g. MOSFETs, IGBTs, and thyristors, as switches to once convert an input DC voltage into an AC voltage by repeated cycles of ON/OFF states of the switches. The converted AC voltage is further converted into a steady DC voltage through the sequence of a transformer, a rectifying circuit, and a smoothing circuit, and the steady DC voltage is then output. In the switching power supply, a transfer factor, i.e. the ratio of the output voltage to the input voltage, is substantially determined by the duty factor of the switches. Accordingly, the switching power supply controls the duty factor of the switches by the control over the switches, and thereby stabilizes the output DC voltage.
Since a switching loss, i.e. the power loss caused by switching, is generally small, the switching power supply can supply power at high efficiency. Hence, the switching power supply is excellent for energy saving.
The transfer factor of the switching power supply substantially depends on only the duty factor of the switches, but does not substantially depend on the switching frequency, i.e. the frequency of the turning ON/OFF of the switches. Furthermore, in the switching power supply, when the switching frequency is raised higher, reactive elements, such as transformers, inductors, and capacitors, is miniaturized with the respective performances of the elements preserved. Therefore, the switching power supply is miniaturized with relative ease, maintaining a steady output voltage.
The following is the description of a switching power supply performing hard switching that is an example of a conventional switching power supply. FIG. 6 is the circuit diagram of the conventional switching power supply 100.
In the switching power supply 100, four switching sections 101H, 101L, 102H, and 102L, form a full-bridge on the primary side of a transformer 3. The switching sections 101H and 102H connected to a high-potential input terminal 1a of the full-bridge are referred to as high-side switching sections. The switching sections 101L and 102L connected to a low-potential input terminal 1b of the full-bridge are referred to as low-side switching sections. The four switching sections include respective switching devices 1HS, 1LS, 2HS and 2LS. These switching devices are semiconductor devices, for example, IGBTs. The four switching sections include respective parasitic capacitors 1HC, 1LC,2HC and 2LC, in parallel with the switching devices. A switching control circuit 70 controls the turning-ON/OFF of the four switching sections.
FIG. 7 is a diagram showing the waveforms of the currents and voltages occurring at various sections of the circuit shown in FIG. 6, according to the hard switching by the switching control circuit 70. In this diagram, positive directions of the currents and voltages at the various sections of the circuit are defined as.
The switching control circuit 70 outputs switching signals G1, G2, G3, and G4 to the switching devices 1HS, 1LS, 2HS, and 2LS, respectively. The switching signals G1, G2, G3, and G4 are rectangular waves. Each switching device is ON during the interval that the switching signal corresponding thereto stays high (H), and OFF during the interval that the switching signal stays low (L).
The switching control circuit 70 performs hard switching for the turning ON/OFF of the four switching sections. Here, hard switching refers to the switching for the simultaneous turning-ON/OFF between one of the high-side switching sections and one of the low-side switching sections. In the hard switching of the switching control circuit 70, the following three periods are sequentially achieved with predetermined time lengths and predetermined cycle periods: (1) A first period corresponds to the period T0-T1 in FIG. 7. During the first period, the first high-side switching section 101H and the second low-side switching section 102L are ON, and the second high-side switching section 102H and the first low-side switching section 101L are OFF. (2) A second period corresponds to the period T2-T3 in FIG. 7. During the second period, the first high-side switching section 101H and the second low-side switching section 102L are OFF, and the second high-side switching section 102H and the first low-side switching section 101L are ON. (3) A third period corresponds to each of the period T1-T2 and the period T3-T4, and is achieved during intervals between the first period and the second period. During the third period, all the four switching sections are OFF.
The following is the description of the hard switching of the switching control circuit 70 in the time sequence from the time T0 to the time T4 shown in FIG. 7.
 less than Period T0-T1 greater than 
At the time T0, the switching control circuit 70 simultaneously changes the first switching signal G1 and the fourth switching signal G4 from L to H, thereby turning ON the first high-side switching section 101H and the second low-side switching section 102L. On the other hand, the first low-side switching section 101L and the second high-side switching section 102H are both OFF.
During the period T0-T1, a substantially steady and positive input voltage Vin is applied across the primary winding 3a of the transformer 3 via the first high-side switching section 101H and the second low-side switching section 102L. Thus, a primary voltage Vt, i.e. the voltage across the primary winding 3a, is substantially equal to the input voltage Vin. Furthermore, the primary current It of the transformer 3 flows from the first junction point P to the second junction point Q of the primary winding 3a. In other words, the primary current It flows in the direction of the arrow shown in FIG. 6. Then, a positive voltage Vin/n is induced across each of the first secondary winding 3b and the second secondary winding 3c of the transformer 3.
Here, the turn ratio of the primary winding 3a, the first secondary winding 3b, and the second secondary winding 3c of the transformer 3 is n:1:1, where n is a positive real number. Since a first rectifying diode 4b is ON, the voltage V5 across a smoothing inductor 5 is substantially equal to Vin/nxe2x88x92Vout. Here, an output voltage Vout, i.e. the voltage across a smoothing capacitor 6, is positive. The output voltage Vout may be assumed to be substantially steady, since the smoothing capacitor 6 has a sufficiently large capacitance. Accordingly, the current I5 flowing through the smoothing inductor 5 increases linearly in the direction of the arrow indicated in FIG. 6 during the period T0-T1. Note that the current 15 increases slowly, since the inductance of the smoothing inductor 5 is sufficiently large. The voltage Vc across a second rectifying diode 4c is substantially equal to +2Vin/n, where the positive direction of the voltage is defined as the direction of the arrow shown in FIG. 6, i.e. the direction of the reverse bias applied to the diode. Accordingly, the second rectifying diode 4c is OFF. Therefore, the current 15 of the smoothing inductor 5 is substantially equal to the current Ib flowing through the first rectifying diode 4b. As a result, during the period T0-T1, the secondary current of the transformer 3 flows only through the first secondary winding 3b, and increases linearly.
The primary current It of the transformer 3 is equal to the sum of the exciting current for the transformer 3 and the equivalent primary current depending on the secondary current of the transformer 3. As shown in FIG. 7, during the period T0-T1, the primary voltage Vt is maintained at the substantially steady value Vin. Accordingly, the exciting current in the primary current It linearly increases in substance. On the other hand, the equivalent primary current increases linearly, since the secondary current of the transformer 3 increases linearly as described above. As a result, the primary current It linearly increases in substance.
 less than Period T1-T2 greater than 
At the time T1, the switching control circuit 70 simultaneously changes the first switching signal G1 and the fourth switching signal G4 from H to L, thereby turning OFF the first high-side switching section 101H and the second low-side switching section 102L. All the four switching sections are thus OFF. Accordingly, the input voltage Vin is not applied across the primary winding 3a of the transformer 3. In other words, the primary voltage Vt of the transformer 3 abruptly drops from the input voltage Vin to zero. Therefore, the respective induced voltages across the secondary windings 3b and 3c of the transformer 3 abruptly change to zero. Then, the voltage Vc across the second rectifying diode 4c abruptly changes to zero, since the first rectifying diode 4b is already ON. Accordingly, the second rectifying diode 4c is abruptly turned ON. As a result, the secondary current of the transformer 3 begins to flow through both of the first secondary winding 3b and the second secondary winding 3c. Thus, at the time T1, commutation occurs on the secondary side of the transformer 3.
During the period T1-T2, no voltage is applied across the primary winding 3a of the transformer 3, and the primary voltage Vt is thereby zero. Thus, no voltage is induced across the secondary windings 3b and 3c of the transformer 3. The voltage V5 across the smoothing inductor 5 is substantially equal to a negative steady voltage xe2x88x92Vout. As a result, the current I5 of the smoothing inductor 5 decreases linearly with a slight inclination.
At the time T1, the primary current It of the transformer 3 abruptly decreases to zero, since all the four switching sections are OFF. Accordingly, during the period T1-T2, the secondary current of the transformer 3 flows through each of the two secondary windings 3b and 3c, wherein one part of the secondary current flowing through the first secondary winding 3b is substantially equal in amount and opposite in direction to the other part flowing through the second secondary winding 3c, so that the equivalent primary current is zero.
 less than Period T2-T3 greater than 
At the time T2, the switching control circuit 70 simultaneously changes the second switching signal G2 and the third switching signal G3 from L to H, thereby turning ON the first low-side switching section 101L and the second high-side switching section 102H. On the other hand, the first high-side switching section 101H and the second low-side switching section 102L are both OFF.
The input voltage Vin is applied across the primary winding 3a of the transformer 3 in the opposite direction to that during the period T0-T1, when the first low-side switching section 101L and the second high-side switching section 102H are ON. In other words, the primary voltage Vt of the transformer 3 abruptly drops from zero to xe2x88x92Vin. Accordingly, the induced voltage across each of the secondary windings 3b and 3c of the transformer 3 abruptly drop from zero to xe2x88x92Vin/n. Then, the second rectifying diode 4c is already ON. Accordingly, the voltage Vb across the first rectifying diode 4b is abruptly raised to +2Vin/n, where the positive direction of the voltage is defined as the direction of the arrow shown in FIG. 6, i.e. the direction of the reverse bias applied to the diode. Thereby, the first rectifying diode 4b is abruptly turned OFF. As a result, the secondary current of the transformer 3 begins to flow only through the second secondary winding 3c. Thus, at the time T2, commutation occurs on the secondary side of the transformer 3.
During the period T2-T3, the first low-side switching section 101L and the second high-side switching section 102H are ON. Accordingly, the primary voltage Vt is substantially equal to xe2x88x92Vin. Furthermore, the primary current It of the transformer 3 flows from the second junction point Q to the first junction point P of the primary winding 3a in the opposite direction to that during the period T0-T1. Then, the voltage across each of the first secondary winding 3b and the second secondary winding 3c is substantially equal to xe2x88x92Vin/n. Since the first rectifying diode is OFF and the second rectifying diode is ON, the voltage V5 across the smoothing inductor 5 is substantially equal to Vcxe2x88x92Vout=Vin/nxe2x88x92Vout. Accordingly, the current I5 flowing through the smoothing inductor 5 linearly increases in the direction of the arrow shown in FIG. 6, in the similar manner to that during the period T0-T1. Since the first rectifying diode 4b is OFF, the current I5 of the smoothing inductor 5 is substantially equal to the current Ic flowing through the second rectifying diode 4c. In other words, during the period T2-T3, the secondary current of the transformer 3 flows only through the second secondary winding 3c and increases linearly.
As shown in FIG. 7, during the period T2-T3, the primary voltage Vt is maintained at the substantially steady value xe2x88x92Vin, in the similar manner to that during the period T0-T1. Accordingly, the exciting current in the primary current It linearly increases in substance. Here, the direction of the primary current It is opposite to that during the period T0-T1. On the other hand, the equivalent primary current increases linearly, since the secondary current of the transformer 3 increases linearly as described above. As a result, the primary current It linearly increases in substance.
 less than Period T3-T4 greater than 
At the time T3, the switching control circuit 70 simultaneously changes the second switching signal G2 and the third switching signal G3 from H to L, thereby turning OFF the first low-side switching section 101L and the second high-side switching section 102H. Since all the four switching sections are thus OFF, the input voltage Vin is not applied across the primary winding 3a of the transformer 3. In other words, the primary voltage Vt of the transformer 3 is abruptly raised from the input voltage xe2x88x92Vin to zero. Therefore, the induced voltage across each of the secondary windings 3b and 3c of the transformer 3 abruptly changes to zero. Then, the voltage Vb across the first rectifying diode 4b abruptly changes to zero, since the second rectifying diode 4c is already ON. Thereby, the first rectifying diode 4b is abruptly turned ON. As a result, the secondary current of the transformer 3 begins to flow through both of the first secondary winding 3b and the second secondary winding 3c, in-the similar manner to that at the time T1. Thus, at the time T3, commutation occurs on the secondary side of the transformer 3.
During the period T3-T4, no voltage is applied across the primary winding 3a of the transformer 3, and the primary voltage Vt is zero. Accordingly, the voltage V5 across the smoothing inductor 5 is substantially equal to the negative steady voltage xe2x88x92Vout. As a result, the current I5 of the smoothing inductor 5 decreases linearly with a slight inclination.
Since all the four switching sections are OFF at the time T3, the primary current It of the transformer 3 abruptly decreases to zero. Accordingly, during the period T3-T4, the secondary current of the transformer 3 flows through each of the two secondary windings 3b and 3c, wherein one part of the secondary current flowing through the first secondary winding 3b is substantially equal in amount and opposite in direction to the other part flowing through the second secondary winding 3c, so that the equivalent primary current is zero, in the similar manner to that during the period T1-T2.
Thus, the state immediately before the time T0 reoccurs in the period T3-T4. After that, the above-described operations during the period T0-T4 are repeated in cycle.
In the hard switching of the switching control circuit 70, the transfer factor, i.e. the ratio of the input voltage Vin to the output voltage Vout, is determined as follows:
Let Ton to be the sum of the time lengths of the first period T0-T1 during which the first high-side switching section 101H is ON, and the second period T2-T3 during which the second high-side switching section 102H is ON. Since the voltage (Vin/nxe2x88x92Vout) is applied across the smoothing inductor 5 during the first and second periods, the magnetic flux to be stored in the smoothing inductor 5 increases by (Vin/nxe2x88x92Vout)xc3x97Ton.
On the other hand, let Toff to be the sum of the time lengths of the period T0-T1 and the period T2-T3, during which all the switching sections are OFF. Since the voltage (xe2x88x92Vout) is applied across the smoothing inductor 5 during each period, the magnetic flux to be stored in the smoothing inductor 5 decreases by Voutxc3x97Toff.
Accordingly, the reset condition of the smoothing inductor 5, i.e. the condition that the increment and decrement of the magnetic flux are balanced in the smoothing inductor 5, is represented by the following equation (1),
(Vin/nxe2x88x92Vout)xc3x97Ton=Voutxc3x97Toff.xe2x80x83xe2x80x83(1)
The transfer factor, i.e. the ratio between the input voltage Vin and the output voltage Vout, is obtained from the equation (1) and represented by the following equation (2),
Vout/Vin=xcex4/n,
where
xcex4=Ton/(Ton+Toff).xe2x80x83xe2x80x83(2)
As the equation (2) indicates, the control over the duty factor xcex4 for the ON/OFF times of the high-side switching sections stably maintains the output voltage Vout at a substantially steady value.
However, the hard switching of the switching control circuit 70 has the following disadvantages: Each switching section of the full-bridge includes the parasitic capacitor connected in parallel with its switching device as shown in FIG. 6. When all the switching sections are OFF, each parasitic capacitor becomes stable in a charged state. In the hard switching, one high-side switching section and one low-side switching section are simultaneously turned ON. Accordingly, when each switching section is turned ON, its switching device connected in parallel with the parasitic capacitor is turned ON under the condition that the parasitic capacitor is charged to a certain extent. Then, the parasitic capacitor is short-circuited via the switching device under the ON condition, and the parasitic capacitor discharges abruptly. Thereby, a surge current occurs in the switching section and is converted into heat or an electromagnetic wave. The electric power is dissipated outside through the heat or the electromagnetic wave. Thus, the switching loss, i.e. the power loss caused by the switching, increases. Furthermore, the heat being due to the surge current causes fatigue in the switching section, and the electromagnetic wave produces noise to peripheral circuit devices.
For example, immediately before the time T0 in FIG. 7, the voltage V1H across the first high-side switching section 101H and the voltage V2L across the second low-side switching section 102L are each maintained at finite values. Accordingly, the parasitic capacitor 1HC of the first high-side switching section 101H and the parasitic capacitor 2LC of the second low-side switching section 102L are charged in proportion to the voltages V1H and V2L, respectively. When both the switching sections are turned ON at the time T0, both of the parasitic capacitors 1HC and 2LC discharge abruptly, and surge currents occur. Thereby, the current I1H flowing through the first high-side switching section 101H and the current I2L flowing through the second low-side switching section 102L increase so as to reach steep peaks sc. When the first low-side switching section 101L and the second high-side switching section 102H are simultaneously turned ON at the time T2, the currents I1L and I2H flowing therethrough have similar peaks.
In the hard switching, furthermore, one high-side switching section and one low-side switching section are simultaneously turned OFF. Accordingly, when the switching sections are OFF, the current supplied to the primary winding of the transformer decreases abruptly. Then, resonance occurs between the leak inductance of the primary winding and the parasitic capacitors in the switching sections, thereby causing surge voltages. The occurrence of the surge voltages causes, for example, energy storage and dissipation by the leak inductance of the primary winding. Thus, the switching loss increases.
For example, in FIG. 7, immediately before the time T1 at which the first high-side switching section 101H and the second low-side switching section 102L are turned OFF, the substantially equal currents I1H=I2L flow through the switching sections, respectively. At the time T1 when both the switching sections are turned OFF, intense resonance occurs between the respective parasitic capacitor 1HC and 2LC and the leak inductance of the primary winding 3a of the transformer 3, and then surge voltages occur. Thereby, the voltage V1H across the first high-side switching section 101H and the voltage V2L across the second low-side switching section 102L change to reach steep peaks sv as shown in FIG. 7. Similar peaks appear on the voltage V1L across the first low-side switching section 101L and the voltage V2H across the second high-side switching section 102H at the time T3 when these switching sections are simultaneously turned ON.
As the above-described, in the hard switching, the surge currents/voltages occur each time of the turning-ON/OFF of the switching sections, thereby increasing the switching loss. The increase of the switching loss is undesirable, since it reduces the energy efficiency of the switching power supply. Furthermore, the above-described surge currents/voltages cause electromagnetic waves at the switching frequency and the harmonics thereof. These electromagnetic waves are undesirable, since they hinder the operations of surrounding devices.
In recent years, there is an intensely growing demand for energy-saving and miniaturization of various electronic equipments. That leads to an intense demand for improvements in efficiency, miniaturization, and output stability of switching power supply. The switching frequency must be further raised in order to satisfy such demands. However, the higher the switching frequency is raised, the larger the switching loss increases. Hence, further raising the switching frequency requires a switching technology capable of reducing the switching loss. Soft switching is known as such a switching technology. Here, soft switching means the following switching: (1) causing resonance to occur between the parasitic capacitor in a switch and an external inductor during the transition from ON to OFF or from OFF to ON; (2) switching from ON to OFF or from OFF to ON when the resonance voltage or the resonance current is zero. In particular, switching with the zero voltage across the switch is referred to as zero volt switching (ZVS).
According to the soft switching, no electric power is dissipated in a switch at the time of the turning-ON/OFF. Hence, no switching loss occurs in principle. According to the ZVS, in particular, no charge remains in the parasitic capacitor of the switch at the time of the turning-ON. Therefore, no surge current occurs.
A conventional switching power supply 110 achieving the ZVS is, for example, disclosed in Japanese Laid-open Patent Application No. Hei 11-89232. FIG. 8 shows the circuit diagram of the switching power supply. The same reference signs as those in FIG. 6 designate similar components to those of the conventional example 100 shown in FIG. 6. This switching power supply 110 differs from the above-described conventional example 100 at the following two points: (1) The switching sections 1H, 1L, 2H, and 2L in the full-bridge include diodes 1HD, 1LD, 2HD, and 2LD, respectively, together with the switching devices and the parasitic capacitors. The diodes are connected in parallel with the respective switching devices. The cathode of the diode is thus connected to the high potential side of the switching device, and the anode of the diode is connected with the low potential side of the switching device. When the switching device is a transistor, such as an IGBT, the diode may be the body diode of the transistor.
(2) The switching control circuit 7 performs soft switching for the switching sections in the full-bridge. Hence, the switching loss reduces in comparison with the above-described conventional example 100. The following is the description of the soft switching of the switching control circuit 7.
FIG. 9 is a diagram showing the waveforms of the currents and voltages occurring at various sections of the circuit shown in FIG. 8, according to the soft switching by the switching control circuit 7. In this diagram, the positive directions of the currents and voltages at the various sections of the circuit are defined as the directions of the arrows in FIG. 8.
The switching control circuit 7 outputs switching signals G1, G2, G3, and G4 to the switching devices 1HS, 1LS, 2HS, and 2LS, respectively. The switching signals G1, G2, G3, and G4 are rectangular waves. Each switching device is ON during the interval that the switching signal corresponding thereto stays high (H), and OFF during the interval that the switching signal corresponding thereto stays low (L).
In the soft switching of the switching control circuit 7, the following four periods are sequentially achieved with predetermined time lengths and predetermined cycle periods: (1) A first period corresponds to the period T0-T1 in FIG. 9. During the first period, the first high-side switching section 1H and the second low-side switching section 2L are ON, and the second high-side switching section 2H and the first low-side switching section 1L are OFF. (2) A second period corresponds to the period T4-T5 in FIG. 9. During the second period, the first high-side switching section 1H and the second low-side switching section 2L are OFF, and the second high-side switching section 2H and the first low-side switching section 1L are ON. (3) A third period corresponds to each of the period T2-T3 and the period T6-T7, and is achieved during intervals between the first period and the second period. During the third period, the first high-side switching section 1H and the second high-side switching section 2H are OFF, and the first low-side switching section 1L and the second low-side switching section 2L are ON. (4) A dead time is an infinitesimal period inserted between two of the above-described three periods, and corresponds to each of the periods T1-T2, T3-T4, T5-T6, and T7-T8. During the dead time, either of the pair of the first high-side switching section 1H and the first low-side switching section 1L, or the pair of the second high-side switching section 2H and the second low-side switching section 2L is OFF. In other words, only one of the four switching sections is ON, and three other switching sections are OFF.
The following is the description of the soft switching of the switching control circuit 7 in the time sequence from the time T0 to the time T8 shown in FIG. 9.
 less than Period T0-T1 greater than 
During the period T0-T1, both of the primary current It of the transformer 3 and the current I5 flowing through the smoothing inductor 5 increase linearly, in a similar manner to that during the period T0-T1 of the above-described conventional example. Then, the secondary current of the transformer 3 flows only through the first secondary winding 3b. 
 less than Period T1-T2 greater than 
At the time T1, the switching control circuit 7 changes the first switching signal G1 from H to L, thereby turning OFF the first high-side switching section 1H. On the other hand, the second low-side switching section 2L is maintained ON. Then, resonance occurs among the leak inductance of the primary winding 3a, the parasitic capacitor 1HC of the first high-side switching section 1H, and the parasitic capacitor 1LC of the first low-side switching section 1L. Because of the resonance, the primary current It simultaneously causes the parasitic capacitor 1HC of the first high-side switching section 1H to charge, and the parasitic capacitor 1LC of the first low-side switching section 1L to discharge. Accordingly, the voltage V1H across the first high-side switching section 1H is smoothly raised from zero, and the voltage V1L across the first low-side switching section 1L drops smoothly from the maximum value Vin.
Immediately before the time T2, the voltage V1H across the first high-side switching section 1H reaches the maximum value Vin. At the same time, the voltage V1L across the first low-side switching section 1L reaches zero. Then, the diode 1LD of the first low-side switching section 1L is turned ON, thereby clamping the voltage V1L at zero. At the time T2, the switching control circuit 7 changes the second switching signal G2 from L to H, thereby turning ON the first low-side switching section 1L. Thus, ZVS is achieved for the turning-ON of the first low-side switching section 1L.
 less than Period T2-T3 greater than 
During the period T2-T3, the primary winding 3a is short-circuited via the two low-side switching sections 1L and 2L. During this period, the primary voltage Vt is substantially zero. Hence, commutation occurs on the secondary side of the transformer 3, in the similar operation to the above-described operation at the time T1 of the conventional example 100. In other words, the secondary current flows through both of the two secondary windings 3b and 3c, since the two rectifying diodes 4b and 4c are both turned ON. As a result, only a substantially steady voltage (xe2x88x92Vout) is applied to the smoothing inductor 5. Hence, the current I5 flowing through the smoothing inductor 5, and then the secondary current, decreases linearly.
The secondary current flowing through the first secondary winding 3b is larger than that flowing through the second secondary winding 3c during the period T2-T3, in contrast to the above-described conventional example 100, because of the following reasons: Since the voltage Vc across the second rectifying diode 4c drops to zero in the period T1-T2, the second rectifying diode 4c is turned ON at the time T2. However, because of the leak inductance on the secondary side of the transformer 3, the current Ib flowing through the first rectifying diode 4b decreases slowly, and the current Ic flowing through the second rectifying diode 4c increases slowly. Hence, during the period T2-T3, the major portion of the secondary current flows through the first secondary winding 3b, and only the minor remaining portion thereof flows through the second secondary winding 3c. As a result, the equivalent primary current does not decrease very much in the period T2-T3 because of the insufficient cancellation between the secondary windings 3b and 3c, but decreases linearly with a slight inclination by the above-described leak inductance. On the other hand, the exciting current of the primary winding 3a is maintained at a substantially steady value, since the primary voltage Vt is substantially zero. As a result, the primary current It decreases linearly.
 less than Period T3-T4 greater than 
At the time T3, the switching control circuit 7 changes the fourth switching signal G4 from H to L, thereby turning OFF the second low-side switching section 2L. On the other hand, the first low-side switching section 1L is maintained ON. Then, resonance occurs among the leak inductance of the primary winding 3a, the parasitic capacitor 2HC of the second high-side switching section 2H, and the parasitic capacitor 2LC of the second low-side switching section 2L. Because of the resonance, the primary current It simultaneously causes the parasitic capacitor 2HC of the second high-side switching section 2H to discharge, and the parasitic capacitor 2LC of the second low-side switching section 2L to charge. Accordingly, the voltage V2H across the second high-side switching section 2H drops smoothly from the maximum value Vin, and the voltage V2L across the second low-side switching section 2L is smoothly raised from zero.
Immediately before the time T4, the voltage V2L across the second low-side switching section 2L reaches the maximum value Vin. At the same time, the voltage V2H across the second high-side switching section 2H reaches zero. Then, the diode 2HD of the second high-side switching section 2H is turned ON, thereby clamping the voltage V2H at zero. At the time T4, the switching control circuit 7 changes the third switching signal G3 from L to H, thereby turning ON the second high-side switching section 2H. As a result, ZVS is achieved for the turning-ON of the second high-side switching section 2H.
 less than Period T4-T5 greater than 
During the period T4-T5, both of the primary current It of the transformer 3 and the current I5 flowing through the smoothing inductor 5 increase linearly, in the similar manner to that during the second period T2-T3 of the above-described conventional example. However, these currents are opposite in direction to those during the period T0-T1. Furthermore, the secondary current of the transformer 3 flows only through the second secondary winding 3c. 
 less than Period T5-T6 greater than 
At the time T5, the switching control circuit 7 changes the third switching signal G3 from H to L, thereby turning OFF the second high-side switching section 2H. On the other hand, the first low-side switching section 1L is maintained ON. Then, resonance occurs among the leak inductance of the primary winding 3a, the parasitic capacitor 2HC of the second high-side switching section 2H, and the parasitic capacitor 2LC of the second low-side switching section 2L. Because of the resonance, the primary current It simultaneously causes the parasitic capacitor 2HC of the second high-side switching section 2H to charge, and the parasitic capacitor 2LC of the second low-side switching section 2L to discharge. Accordingly, the voltage V2H across the second high-side switching section 2H is raised smoothly from zero, and the voltage V2L across the second low-side switching section 2L drops smoothly from the maximum value Vin.
Immediately before the time T6, the voltage V2H across the second high-side switching section 2H reaches the maximum value Vin. At the same time, the voltage V2L across the second low-side switching section 2L reaches zero. Then, the diode 2LD of the second low-side switching section 2L is turned ON, thereby clamping the voltage V2L at zero. At the time T6, the switching control circuit 7 changes the fourth switching signal G4 from L to H, thereby turning ON the second low-side switching section 2L. As a result, ZVS is achieved for the turning-ON of the second low-side switching section 2L.
 less than Period T6-T7 greater than 
During the period T6-T7, the primary winding 3a is short-circuited via the two low-side switching sections 1L and 2L again. Accordingly, commutation occurs on the secondary side of the transformer 3, in the similar operation to that during the period T2-T3. The secondary current flows through both of the two secondary windings 3b and 3c. However, the secondary current flowing through the second secondary winding 3c is larger than that flowing through the first secondary winding 3b, in the opposite manner to that during the period T2-T3. Then, the equivalent primary current does not decrease very much. Furthermore, only the substantially steady voltage (xe2x88x92Vout) is applied to the smoothing inductor 5. Accordingly, both of the primary current It and the secondary current I5 decrease linearly.
 less than Period T7-T8 greater than 
At the time T7, the switching control circuit 7 changes the second switching signal G2 from H to L, thereby turning OFF the first low-side switching section 1L. On the other hand, the second low-side switching section 2L is maintained ON. Then, resonance occurs among the leak inductance of the primary winding 3a, the parasitic capacitor 1HC of the first high-side switching section 1H, and the parasitic capacitor 1LC of the first low-side switching section 1L. Because of the resonance, the primary current It simultaneously causes the parasitic capacitor 1HC of the first high-side switching section 1H to discharge, and the parasitic capacitor 1LC of the first low-side switching section 1L to charge. Hence, the voltage V1H across the first high-side switching section 1H drops smoothly from the maximum value Vin, and the voltage V1L across the first low-side switching section 1L is smoothly raised from zero.
Immediately before the time T8, the voltage V1L across the first low-side switching section 1L reaches the maximum value Vin. At the same time, the voltage V1H across the first high-side switching section 1H reaches zero. Then, the diode 1HD of the first high-side switching section 1H is turned ON, thereby clamping the voltage V1H at zero. At the time T8, the switching control circuit 7 changes the first switching signal G1 from L to H, thereby turning ON the first high-side switching section 1H. As a result, ZVS is achieved for the turning-ON of the first high-side switching section 1H.
At the time T8, the same state as the time T0 reoccurs. Thus, the operations during the period from the time T0 to the time T8 are repeated in cycle.
The time length of the dead time, i.e. each time length of the period T1-T2, the period T3-T4, the period T5-T6, and the period T7-T8, is sufficiently shorter than any time length of the first period T0-T1, the second period T4-T5, and the third periods T2-T3 and T6-T7. Generally, each time length of the first, the second, and the third periods is about several microseconds, but the time length of the dead time is about several tens to several hundreds of nanoseconds.
When the dead times are ignored in comparison with the first period and other periods, the transfer factor (the ratio between the input voltage Vin and the output voltage Vout) in the above-described soft switching is obtained as follows:
Let Ton to be the sum of the time lengths of the first period T0-T1 and the second period T4-T5. In each of these periods, the voltage (Vin/nxe2x88x92Vout) is applied across the smoothing inductor 5 in the above-described manner. Therefore, the magnetic flux to be stored in the smoothing inductor 5 increases by (Vin/nxe2x88x92Vout)xc3x97Ton in total during the above-described two periods. On the other hand, let Toff to be the sum of the time lengths of the third periods T1-T4 and T5-T8. During each of the third periods, the voltage (xe2x88x92Vout) is applied across the smoothing inductor 5 in the above-described manner. Therefore, the magnetic flux to be stored in the smoothing inductor 5 decreases by Voutxc3x97Toff in total. Accordingly, the reset condition of the smoothing inductor 5 is represented by the similar equation (1) to that of the hard switching:
(Vin/nxe2x88x92Vout)xc3x97Ton=Voutxc3x97Toff.xe2x80x83xe2x80x83(1)
Therefore, the transfer factor is represented by the similar equation (2) to that in the hard switching:
Vout/Vin=xcex4/n,
where
xcex4=Ton/(Ton+Toff).xe2x80x83xe2x80x83(2)
In other words, in the soft switching, in the similar manner to that of the hard switching, the control over the duty factor xcex4 for the ON/OFF time of the switches in the high-side switching sections 1H and 2H stably maintains the output voltage Vout at a substantially steady value.
Furthermore, in the soft switching, the four switching sections 1H, 1L, 2H, and 2L are turned ON according to the above-described ZVS. Hence, the surge currents/voltages caused by the switching do not occur in the switching sections, in contrast to the hard switching. Thus, the switching loss in the soft switching is less than that in the hard switching.
The soft switching has the advantage in switching loss over the hard switching, but is at a disadvantage in respect of conduction loss compared with the hard switching as follows: The soft switching achieves the third periods, which corresponds to the periods T2-T3 and T6-T7 in FIG. 9, i.e. the periods during the short-circuited of the primary winding 3a of the transformer 3. During the third periods, the primary current It circulates through the two low-side switching sections 1L and 2L, and the primary winding 3a. In other words, the primary current It is not zero but finite, during the third periods. In this respect, the soft switching is contrast to the hard switching. Each of the switching sections generally includes an ON resistance, and the primary winding 3a of the transformer 3 generally includes a parasitic resistor. These resistors produce Joule heat and thereby dissipate energy when the primary current It flows. Thus, the conduction loss caused by the primary current It during the third periods (hereafter referred to as circulating current loss) in the soft switching is larger than that in the hard switching.
According to the demand for the miniaturization of switching power supplies, the sizes of the switching sections are restricted. As a result, the ON resistance of the switching section has a lower limit. In addition, the smaller the size of the switching power supply, the larger the lower limit of the ON resistance. In such a miniature switching power supply, the increase of the circulating current loss may be larger than the reduction of the switching loss in the soft switching. Then, the efficiency of the switching power supply does not increase sufficiently.
In the full-bridge type converter based on the above-described soft switching (hereafter referred to as an active clamp full-bridge (ACFB) converter), the leak inductance of the primary winding of the transformer is used to cause resonance together with the parasitic capacitors of the switching sections. However, the leak inductance is inherently small. When the leak inductance is too small, a sufficiently large resonance current cannot be obtained, and the parasitic capacitors of the switching sections may be insufficiently charged or discharged. As a result, ZVS cannot be achieved, and the switching loss may increase.
The source of the surge current/voltage caused by the switching is not limited to components in the switching sections. The surge current/voltage may be caused by the turning-OFF of each of the rectifying diodes 4b and 4c at the time of the commutation on the secondary side in both of the conventional switching power supply 100 according to the above-described hard switching and the ACFB converter 110 according to the soft switching. Thereby, the currents Ib and Ic flowing through the respective rectifying diodes 4b and 4c have peaks cf, and the voltages Vb and Vc across the respective rectifying diodes have peaks vf, as shown in FIGS. 7 and 9. The cause of the occurrence of the surge currents/voltages in the rectifying diodes 4b and 4c as follows:
Either of the rectifying diodes 4b and 4c is turned OFF at each start of the first period and the second period. The starts of the first period and the second period correspond to the times T0 and T2 in FIG. 7, and the times T0 and T4 in FIG. 9. A diode generally stores electric charges while it is ON. Thereby, when the diode is turned OFF by an applied reverse bias, the stored charges are discharged as a reverse current. In the above-described switching power supplies 100 and 110, a surge current occurs as the reverse current, when either of the rectifying diodes 4b and 4c is turned OFF caused by the commutation on the secondary side.
The two rectifying diodes 4b and 4c is connected with the two secondary windings 3b and 3c of the transformer 3, respectively. For example, when the first rectifying diode 4b is turned OFF, resonance occurs between the parasitic capacitance of the first rectifying diode 4b and the leak inductance of the first secondary winding 3b based on the above-described reverse current. Thereby, a surge voltage occurs across the first rectifying diode 4b. Similarly, a surge voltage occurs across the second rectifying diode 4c based on the reverse current, when the diode is turned OFF.
The surge currents/voltages caused by the turning-ON/OFF of the above-described rectifying diodes produce noises. The power loss caused by the noise is a part of the switching loss of the switching power supply, and undesirably reduces the efficiency of the switching power supply.
The object of the present invention is to provide a switching power supply that may reduce the circulating current loss in addition to the reduction of the switching loss by the soft switching.
The switching power supply according to the present invention comprises:
a DC-DC converter
(A) including:
(a) four switching sections of a first high-side switching section, a second high-side switching section, a first low-side switching section, and a second low-side switching section, each having (1) a switching device that is turned ON/OFF according to a switching signal from the outside, (2) a diode connected in parallel with the switching device, and (3) a capacitance connected in parallel with the switching device;
(b) a transformer having (1) a primary winding, and (2) a first secondary winding and a second secondary winding connected in series whose Junction point is a common terminal;
(c) a first rectifying section and a second rectifying section, each having (1) a rectifying device and (2) a snubber connected in parallel with the rectifying device, the snubber having a snubber capacitor; and
(d) a smoothing section having a first input terminal and a second input terminal, for smoothing an input supplied via the input terminals, and for outputting the smoothed input;
in the DC-DC converter,
(B) the cathode of the first high-side switching section is connected with the higher potential terminal of a substantially steady direct voltage source, the anode of the first high-side switching section is connected with the cathode of the first low-side switching section, and the anode of the first low-side switching section is connected with the lower potential terminal of the substantially steady direct voltage source;
(C) the cathode of the second high-side switching section is connected with the higher potential terminal of the substantially steady direct voltage source, the anode of the second high-side switching section is connected with the cathode of the second low-side switching section, and the anode of the second low-side switching section is connected with the lower potential terminal of the substantially steady direct voltage source;
(D) one terminal of the primary winding of the transformer is connected with the junction point between the first high-side switching section and the first low-side switching section, and the other terminal of the primary winding is connected with the junction point between the second high-side switching section and the second low-side switching section;
(E) a terminal other than the common terminal of the first secondary winding of the transformer is connected with the anode of the first rectifying section, a terminal other than the common terminal of the second secondary winding is connected with the anode of the second rectifying section, and the common terminal is connected with the first input terminal of the smoothing section; and
(F) the cathode of each of the first rectifying section and the second rectifying section is connected with the second input terminal of the smoothing section; and
a switching control section for
(A) determining a delay time based on a resonance cycle depending on the equivalent capacitance of the snubber in each of the first rectifying section and the second rectifying section, and the leak inductance on the secondary side of the transformer with the primary winding short-circuited; and
(B) outputting the switching signal with a predetermined switching frequency and a predetermined phase to each of the four switching sections, thereby
(a) achieving a first period and a second period sequentially with predetermined time lengths and predetermined cycle periods, (1) during the first period, the first high-side switching section and the second low-side switching section are ON and the second high-side switching section and the first low-side switching section are OFF, and (2) during the second period, the first high-side switching section and the second low-side switching section are OFF and the second high-side switching section and the first low-side switching section are ON;
(b) at the end of the first period, turning OFF one of the first high-side switching section and the second low-side switching section the delay time later than the turning-OFF of the other switching section; and
(c) at the end of the second period, turning OFF one of the second high-side switching section and the first low-side switching section the delay time later than the turning-OFF of the other switching section.
In the above-described switching power supply, the high-side and low-side switching sections that are ON during each of the first period and the second period are both OFF at the time when the delay time elapsed from each end of the periods. Then, the primary current, i.e. the current flowing through the primary winding of the transformer, quickly attenuates as follows.
During the first period, the first rectifying section is OFF, and during the second period, the second rectifying section is OFF. Thereby, full-wave rectification is achieved on the secondary side of the transformer. Then, in the OFF-side rectifying section, the snubber capacitor stores charges by a reverse bias. At each end of the first period and the second period, the induced voltage across each of the secondary windings drops to zero. Then, the snubber capacitor discharges in the OFF-side rectifying section. The discharge current causes a current to start flowing through the secondary winding connected with the rectifying section before the rectifying device is ON in the rectifying section. Based on the current, resonance occurs between the snubber capacitor and the leak inductance on the secondary side of the transformer. Here, the leak inductance on the secondary side of the transformer is defined as the leak inductance equivalently produced between the terminals other than the common terminal of the two secondary windings when the primary winding is short-circuited. In the description, the leak inductance is referred to as the leak inductance on the secondary side of the transformer with the primary winding short-circuited. Accordingly, the commutation of the secondary current of the transformer progresses smoothly and quickly, in contrast to the conventional switching power supply. In other words, substantially equal amounts of the secondary currents begin to flow through both the secondary windings immediately after each of the first period and the second period. As a result, the equivalent primary current is smoothly and quickly cancelled between the secondary currents flowing through the two secondary windings. Hence, the primary current is smoothly and quickly reduced at each end of the first period and the second period. Therefore, surge voltages are reduced when the primary current is cut off the above-described delay time later, and the switching loss caused by the cut-off of the primary current is thus reduced. Furthermore, the circulating current loss caused by the primary current is reduced, since the primary current is cut off between the first period and the second period.
In the above-described switching power supply, the switching control section may:
(a) at the end of the first period, turns OFF the second low-side switching section the delay time later than the turning-OFF of the first high-side switching section: and
(b) at the end of said second period, turns OFF the first low-side switching section the delay time later than the turning-OFF of the second high-side switching section.
In this switching power supply, the primary current of the transformer circulates through the two low-side switching sections and the primary winding until the corresponding delay time elapsed after each end of the first period and the second period.
Alternatively, the switching control section may:
(a) at the end of the first period, turns OFF the first high-side switching section the delay time later than the turning-OFF of the second low-side switching section; and
(b) at the end of the second period, turns OFF the second high-side switching section the delay time later than the turning-OFF of the first low-side switching section.
In this switching power supply, the primary current of the transformer circulates through the two high-side switching sections and the primary winding until the corresponding delay time elapsed after each end of the first period and the second period.
Apart from the above-described, the switching control section may:
(a) at the end of the first period, turns OFF the second low-side switching section the delay time later than the turning-OFF of the first high-side switching section; and
(b) at the end of the second period, turns OFF the second high-side switching section the delay time later than the turning-OFF of the first low-side switching section.
In this switching power supply, the primary current of the transformer circulates through the two low-side switching sections and the primary winding until the delay time elapsed after the end of the first period. On the other hand, the primary current of the transformer circulates through the two high-side switching sections and the primary winding until the delay time elapsed after the end of the second period. The above-described alternating of the current circulating sections equalizes stresses exerted on the switching sections.
In the above-described switching power supply, the switching control section may:
(A) determines a first dead time based on a resonance cycle depending on the equivalent capacitance of the first high-side switching section, the equivalent capacitance of the first low-side switching section, and the leak inductance of the primary winding of the transformer;
(B) determines a second dead time based on a resonance cycle depending on the equivalent capacitance of the second high-side switching section, the equivalent capacitance of the second low-side switching section, and the leak inductance of the primary winding of the transformer; and
(C) at each end of the first period and the second period, achieves one of the periods
(a) when the first high-side switching section and the first low-side switching section are both OFF during the first dead time; and
(b) when the second high-side switching section and the second low-side switching section are both OFF during the second dead time.
For example, at the end of the first period, the first high-side switching section is turned OFF earlier than the second low-side switching section. Then, resonance occurs among the leak inductance of the primary winding of the transformer and the equivalent capacitances of the first high-side switching section and the first low-side switching section. By the resonance, the capacitor in the first high-side switching section charges, and the capacitor in the first low-side switching section discharges. Furthermore, when the capacitor in the first low-side switching section discharges completely, the diode in the same switching section is turned ON, thereby clamping the voltage across the switching section at zero. In this state, the second low-side switching section is turned ON. Hence, in the above-described switching power supply, ZVS is achieved for the turning-ON of the second low-side switching section. Therefore, the switching loss is reduced. When the second low-side switching section is turned OFF earlier at the end of the first period, and even at the end of the second period, ZVS is similarly achieved for the switching section that is turned ON earlier.
The dead time is determined based on the resonance cycle depending on the equivalent capacitance of the high-side switching section, the equivalent capacitance of the low-side switching section, and the leak inductance of the primary winding of the transformer. Preferably, the dead time is substantially equal to xc2xc times the resonance cycle. On the other hand, the delay time is determined based on the resonance cycle depending on both of the equivalent capacitance of the snubber and the leak inductance on the secondary side of the transformer with the primary winding short-circuited. Preferably, the delay time is substantially equal to xc2xc times the resonance cycle. The delay time is mainly adjusted with the capacitance of the snubber capacitor. Therefore, in the above-described switching power supply, the primary current is cut off immediately after the above-described ZVS is achieved. Thus, the circulating current loss is reduced along with the switching loss.
In the above-described switching power supply, the snubber may include a resistor connected in series with the snubber capacitor. At the starts of the first period and the second period, an induced voltage from the primary side is produced across each of the secondary windings of the transformer. Thereby, either of the two rectifying sections being ON is turned OFF. In the rectifying section that is then turned OFF, resonance occurs between the snubber capacitor and the leak inductance on the secondary side of the transformer. The resistor in the snubber attenuates the resonance quickly. Thereby, the ringing caused by the resonance is suppressed, and in addition, the peak of the voltage applied across the rectifying device in parallel with the snubber is reduced.
Furthermore, in the above-described switching power supply, (a) the snubber may include an auxiliary rectifying device connected in parallel with the resistor; and (b) one pair of the anodes and the cathodes of the rectifying device and the auxiliary rectifying device may be connected with each other. At each end of the first period and the second period, the snubber capacitor discharges in the OFF-side rectifying section. Then, the discharge current flows through the auxiliary rectifying device in parallel with the resistor, thereby avoiding the reduction by the resistor. Accordingly, the peak of the resonance current increases in the resonance between the equivalent capacitance of the snubber and the leak inductance on the secondary side of the transformer. As a result, the equivalent primary current is sufficiently and more quickly cancelled at each end of the first period and the second period, and thus the primary current is sufficiently reduced. Accordingly, not only the conduction loss caused by the resistor in the snubber but also the switching and circulating current losses caused by the primary current are reduced. In addition, the heat generation of the resistor in the snubber caused by the discharge current is suppressed. Thereby, increase of the resistance and impairment of the functions of other devices owing to the heat are prevented.
In the above-described switching power supply, the rectifying device and the auxiliary rectifying device included in the snubber are preferably diodes. Alternatively, they may be semiconductor. switching devices, such as IGBTs and MOSFETs. In this case, the switching control section preferably controls the switching of the semiconductor switching devices in synchronization with the switching of the switching sections in the full-bridge on the primary side.
While the novel features of the invention are set forth particularly in the appended claims, the invention, both as to organization and content, will be better understood and appreciated, along with other objects and features thereof, from the following detailed description taken in conjunction with the drawings.